CMOS RF Modeling for GHz Communication IC ’ s

With the advent of submicron technologies, GHz RF circuits can now be realized in a standard CMOS process [1]. A major barrier to the realization of robust commercial CMOS RF components is the lack of adequate models which accurately predict MOSFET device behavior at high frequencies. The conventional microwave table-lookup-based approach requires a large database obtained from numerous device measurements and computationally intense simulations for accurate results. This method becomes prohibitively complex when used to simulate highly integrated CMOS communication systems; hence, a compact model, valid for a broad range of bias conditions and operating frequencies is desirable. BSIM3v3 has been widely accepted as a standard CMOS model for low frequency applications. Recent work has demonstrated the capability of modeling CMOS devices at high frequencies by utilizing a complicated substrate resistance network and extensive modification to the BSIM3v3 source code [2]. This paper first describes a unified device model realized with a lumped resistance network suitable for simulations of both RF and baseband analog circuits; then verifies the accuracy of the model to measured data on both device and circuit levels. The new BSIM3v3 RF model is realized with the addition of three resistors R g , R subd , and R subs to the existing BSIM3v3.1 model (shown in Fig. 1). R g models both the physical gate resistance as well as the non-quasi-static (NQS) effect. R subd and R subs are the lumped substrate resistances between the source/drain junctions and the substrate contacts. The values of R subd and R subs may not be equal as they are functions of the transistor layout (illustrated in Fig. 2). To demonstrate the accuracy of the model, s-parameters of the BSIM3v3 RF model, BSIM3v3 model, and measured data of a 0.35µm NMOS device are plotted in Fig. 3. The improvement can be clearly seen from the S 22 but hardly from S 11. A better picture and more physical insight may be obtained by separating the terminal impedance into the real and imaginary parts with the following six parameters, R in = real(1/ y 11) ; C in = −1/ imag(1/ y 11)/ ω ; R out = 1/ real(y 22) ; C out = imag(y 22)/ ω ; g m = real(y 21) ; C fb = −imag(y 12)/ ω , where ω is the frequency in rad/s (gate is port1, drain port2, and body shorted to the source). As …